1. Field of the Invention
This invention relates to phase shifters and, more particularly, to a phase shifter circuit which provides phase shifting of signals with high linearity at high power levels.
2. Description of Related Art
Altering the phase of a Radio Frequency (RF) signal is often necessary in the processing of RF signals within receivers and transmitters. Many methods have been developed to achieve phase shift of RF signals. Mechanical methods exist which rely on varying the physical length of transmission line to change the electrical length to provide phase shift of the output signal with respect to the input signal. Electrically variable methods exist which rely on controlled reflection of RF signals to alter the phase shift between input and output. One common implementation allows the user to adjust the phase of an RF signal electrically by varying a control voltage. Such circuits have a limitation on the maximum RF signal power level. Operating these circuits beyond a specific RF power level will cause significant non-linear distortion of other RF output signal properties.
FIG. 1 shows a phase shifter circuit 10 based on the use of a 90 degree hybrid coupler 12. A variable capacitive reactance is placed on the 0 degree port 14 by varactor diode CR1 and on the -90 degree port 16 by varactor diode CF2. An input signal applied to the input port 18 is reflected from the 0 degree and -90 degree ports 14 and 16 and appears at an isolated port 20 as the resulting output signal. The phase relationship between input and output signals depends upon the value of the capacitive reactance present on the 0 degree port 14 and -90 degree port 16. The capacitive reactance of a circuit is the impedance of the circuit due to the presence of capacitance. The capacitive reactance X.sub.c of a circuit can be given by 1/.omega.C where .omega. is the frequency of a signal through the circuit and C is the capacitance. The amount of signal loss experienced between input and output is dependent upon how well the capacitance reactance on the 0 degree port 14 is matched to the -90 degree port 16. As such, the reactances are made equal in value to one another. The value of the capacitive reactance is changed by varying the reverse DC bias voltage on diodes CR1 and CR2. As such, the range of phase shift between input and output signals is dependent upon the range of capacitive reactance achieved by changing the DC control voltage V.sub.cntl.
At low RF power levels, the output signal is nearly identical to the input signal in all respects except phase shift. The amplitude of the output signal is slightly lower due to a small but finite loss through the hybrid coupler 12. As the power level of the RF signal increases, the voltage swing of the RF signal begins to effect the DC bias conditions on the diodes CR1 and CR2. The bias voltage on CR1 becomes different than the bias voltage on CR2, resulting in an imbalance between the capacitive reactances on the 0 degree port 14 and the -90 degree port 16. The imbalance of capacitive reactances between the two ports 14 and 16 causes an amplitude distortion or loss of the output signal at the output (isolated) port 20. The effects of this non-linear distortion at higher RF signal voltage levels can be measured with a two tone Inter-Modulation Distortion (IMD) test. Two RF signals having equal amplitudes at different frequencies (f1 and f2) are applied to the input port 18 and monitored with a spectrum analyzer (not shown) at the output (isolated) port 20. At low RF signal power levels, only the two input signals are present at the output port 20 and are shifted in phase with respect to the two input signals at the input port 18. As the power levels of the two input signals are increased, non-linear distortion occurs as described above and causes mixing of the two RF signals. Newsignals appear at the output port at frequencies of (2.times.f2)-f1 and (2.times.f1)-f2. These distortion components are commonly referred to as third-order intermodulation distortion (IMD) products because they are produced by the non-linear multiplication of the two original RF signals. This is an undesirable effect because the IMD products appear as new RF signals which use up frequency spectrum and contain invalid information (a mixture of f1 and f2).
FIG. 2 shows an embodiment of a phase shifter 22 which uses tvo pairs 24 and 26 of varactor diodes connected back to back (common cathode) to provide capacitive reactance on the 0 degree port 14 and the -90 degree port 16, respectively. Using two pairs 24 and 26 of back to back diodes extends the RF signal power level range in which the phase shifter remains linear. FIG. 3a shows a back to back diode arrangement 28 to explain how the diode pairs 24 and 26 are used in the phase shifter 22. RF input signal voltage levels -V.sub.s large enough to cause changes in the DC bias voltage across the diodes D1 and D2 are divided equally across diodes D1 and D2, thereby providing less of a change in the DC bias voltage across the diodes D1 and D2. The positive voltage impressed across D1 functions to decrease the capacitive reactance of D1, and the positive voltage impressed across D2 functions to increase the capacitive reactance of D2. As such, the effect on D1 cancels the eifect on diode D2. Thus, the single back to back diode configuration maintains more stable capacitive reactance values at higher RF input signal voltage levels V.sub.s. FIG. 3b illustrates how the capacitive reactance of diode D1 cancels the effects of diode D2 when the RF signal reverses polarity for the single back to back diode configuration 28. As in FIG. 3a, RF signal voltage levels -V.sub.s large enough to cause changes in the DC bias voltage across the diodes D1 and D2 are divided equally across diodes D1 and D2. The negative voltage impressed across D1 functions to increase the capacitive reactance of D1. The negative voltage impressed across D2 functions to decrease the capacitive reactance of D2. Again, the effect on the change in capacitive reactance on D1 cancels the effect on the change of capacitive reactance on the diode D2.
The phase shifter 22 of FIG. 2 uses matched diodes D1 and D2 for the diode pair 24 and matched diodes D3 and D4 for the diode pair 26. Matched diodes are used in an attempt to prevent imbalances in the capacitive reactance responses between the diodes D1 and D2 for the diode pair 24 and between the diodes D3 and D4 of diode pair 26, but does not prevent imbalances between the pairs 24 and 26. Using matched diodes can also help to reduce parasitic inductances which can effect the capacitive reactances of the diode pairs 24 and 26. An imbalance in the capacitive reactance between the 0 and -90 degree ports 14 and 16 should be avoided to prevent an increase in the insertion loss between the input and the output ports 18 and 20. Differences in the capacitive reactance responses between the diodes D1 and D2 and between the diodes D3 and D4 can create distortion and cause imbalances in the capacitive reactance between the 0 degree port 14 and the -90 degree port 16 of the coupler 12. Diodes available in the same package made from the same wafer generally guarantee similar characteristics. Using the standard two tone IMD test, an improvement of more than 18 decibels (dB) in the difference between the output signal power level and the IMD power level was found between the phase shifter 22 (FIG. 2) and the phase shifter 10 (FIG. 1) for the same input power levels.
Today, high power linear amplifiers operating in the cellular and PCS frequency bands (869-894 MHz and 1930-1990 MHz) produce signals having power levels much greater than typically encountered in the past. These high power level signals test the limits of current phase shifter designs. At these high power levels and even using matched diodes, imbalances in the capacitive reactance between the control ports 14 and 16 can result in insertion losses between the input and output ports 18 and 20. Wide voltage swings in the RF signal can cause such capacitive reactance mismatches due to different capacitive reactance responses between the diode pairs 24 and 26. The differences in the capacitive responses may reflect capacitive reactance response differences between the diodes D1 and D2 and/or between diodes D3 and D4 within a diode pair 24 or 26. Thus, a need exists for a voltage-variable phase shifter circuit with an extended linear power level range which is capable of changing only the phase of RF signals at high power levels.